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Mobile Phone Patent Abstract
According to a second embodiment of the invention, a mobile phone
receiver comprises a first down converter using a first local oscillator
frequency which can be tuned in frequency steps by a programmable
digital frequency synthesizer PLL which is locked to a reference
frequency. The first down converter converts received signals to
a first IF for filtering. A second down converter using a second
local oscillator converts first IF signals to a second IF. The second
local oscillator frequency is generated using a second digital frequency
synthesizer PLL which locks the second oscillator to the reference
frequency. A third down converter mixes the transmit frequency with
the first local oscillator frequency to produce a lock frequency.
A third digital frequency synthesizer PLL compares the lock frequency
and the reference frequency to control generation of the transmit
frequency.
Mobile Phone Patent Claims
What is claimed is:
1. A radio transmitting and receiving apparatus for generating
a signal for transmission on a transmit frequency and for receiving
a signal on a receive frequency, comprising:
a first downconvertor for mixing said received signal with a first
local oscillator frequency signal and converting it to a first intermediate
frequency signal;
a second downconvertor for mixing said first intermediate frequency
signal with a second local oscillator frequency signal and converting
it to a second intermediate frequency;
reference means for providing a reference frequency signal;
first local oscillator frequency synthesizer means having a first
input for said first local oscillator frequency signal and a second
input for said reference frequency signal and in dependence thereon
producing a control signal to control said first local oscillator
to produce the desired first local oscillator frequency signal;
second local oscillator frequency synthesizer means having a first
input for said second local oscillator frequency signal and a second
input for said reference frequency signal and producing in dependence
thereon a control signal to control said second local oscillator
to produce the desired second local oscillator frequency signal;
transmit oscillator means for generating a transmit frequency signal;
transmit downconvertor means for mixing the transmit frequency
signal with the first local oscillator signal to produce a lock
frequency signal;
transmit synthesizer means having a first input for the lock frequency
signal and a second input for the reference frequency signal and
in dependence thereon producing a control signal for the transmit
oscillator means to control the transmit frequency signal.
2. The apparatus of claim 1, further comprising a frequency doubler
for doubling said first local oscillator frequency signal input
to said first local oscillator frequency synthesizer means, wherein
said first local oscillator frequency synthesizer means selects
said first local oscillator frequency signal or the doubled first
oscillator frequency for low band operation and high band operation,
respectively.
3. The apparatus of claim 1, wherein the first down converter mixes
said received signal with the first local oscillator frequency signal
for low band operation and mixes said received signal with a doubled
first oscillator frequency signal for high band operation.
4. The apparatus of claim 1, further comprising a frequency doubler
for doubling the transmit frequency signal for high band operation.
5. The apparatus of claim 1, further comprising a frequency divider
for dividing the transmit frequency signal for low band operation.
6. The apparatus of claim 1, further comprising frequency multipliers
for multiplying the transmit frequency signal for low band operation
and/or high band operation.
7. The apparatus of claim 1, further comprising frequency dividers
for dividing the transmit frequency signal for low band operation
and/or high band operation.
8. The apparatus of claim 1, further comprising a filter for filtering
the lock frequency signal.
9. The apparatus of claim 1, further comprising a modulator for
modulating the transmit frequency signal to produce the signal for
transmission.
10. The apparatus of claim 1, further comprising a variable gain
amplifier for amplifying the transmit frequency signal to produce
the signal for transmission.
11. A method for generating a signal for transmission on a transmit
frequency and for receiving a signal on a receive frequency, comprising
the steps of:
mixing said received signal with a first local oscillator frequency
signal to produce a first mixed signal;
converting said first mixed signal to a first intermediate frequency
signal;
mixing said first intermediate frequency signal with a second local
oscillator frequency signal to produce a second mixed signal;
downconverting said second mixed signal to a second intermediate
frequency;
comparing said first local oscillator frequency signal and a reference
frequency signal and producing in dependence thereon a control signal
to control the production of the desired first local oscillator
frequency signal;
comparing said second local oscillator frequency signal and said
reference frequency signal and producing in dependence thereon a
control signal to control production of the desired second local
oscillator frequency signal;
generating a transmit frequency signal;
mixing the transmit frequency signal with the first local oscillator
signal to produce a lock frequency signal;
comparing the lock frequency signal and the reference frequency
signal and in dependence thereon producing a control signal to control
the generation of the transmit frequency signal.
12. The method of claim 11, further comprising the steps of:
doubling said first local oscillator frequency signal; and
selecting the first local oscillator frequency signal or the doubled
first local oscillator frequency signal for comparison with the
reference frequency signal for low band operation and high band
operation, respectively.
13. The method of claim 11, wherein said received signal is mixed
with the first local oscillator frequency signal for low band operation,
and said received signal is mixed with a doubled first oscillator
frequency signal for high band operation.
14. The method of claim 11, further comprising a step of doubling
the transmit frequency signal for high band operation.
15. The method of claim of claim 11, further comprising a step
of dividing the transmit frequency signal for low band operation.
16. The method of claim 11, further comprising a step of multiplying
the transmit frequency signal for low band operation and/or high
band operation.
17. The method of claim 11, further comprising a step of dividing
the transmit frequency signal for low band operation and/or high
band operation.
18. The method of claim 11, further comprising a step of filtering
the lock frequency signal.
19. The method of claim 11, further comprising a step of modulating
the transmit frequency signal to produce the signal for transmission.
20. The method of claim 11, further comprising a step of amplifying
the transmit frequency signal for transmission.
Mobile Phone Patent Description
FIELD OF THE INVENTION
The invention relates to transceiver architecture in a mobile phone.
More particularly, this invention relates to on-channel transceiver
architecture in a dual band mobile phone.
BACKGROUND OF THE INVENTION
It is well known in the art of mobile radiotelephones to employ
a receiver for receiving a receive frequency signal while a transmitter
simultaneously transmits a transmit frequency signal in the other
direction, the transmit frequency being separated from the receive
frequency by a constant offset known as the duplex spacing.
Although the duplex spacing is nominally a constant, it can be
a different constant depending on the frequency band in which the
mobile phone is operating. Complications can then arise in constructing
mobile phones that operate in more than one frequency band. Various
approaches have been proposed for handling these complications.
One approach is to employ two separate and independent receiver-transmitter
chains for each band of interest. This has the advantage of inherent
redundancy for enhanced operation reliability and power control.
However, the separate receiver-transmitter chains require extra
space, resulting in a more expensive and bulkier mobile phone product.
As the market demand shifts toward smaller, less expensive phones
with more embedded features, this approach becomes less suitable.
U.S. patent application Ser. No. 08/795,930 entitled "Transmit
Signal Generation with the Aid of Receiver" (Dolman) describes
the use of the second local oscillator of the receiver as a reference
frequency against which a transmit frequency is controlled relative
to a receive frequency to achieve either a first or a second duplex
spacing. The Dolman application is hereby incorporated by reference.
It is also known in the prior art, when packaging two synthesizer
PLL circuits into a common integrated circuit, to synchronize or
otherwise relate the reference dividers of the two PLLs so that
their phase comparators do not mutually interfere. The Philips UM1005
and 8026 dual synthesizer integrated circuits available on the open
market use this technique. These circuits include the use of fractional-N
dividers and programmable loop bandwidth, such as described in U.S.
Pat. Nos. 5,095,288 and 5,180,993, which are hereby incorporated
by reference. Novel ways to employ such synthesizers in dual mode
satellite/cellular telephones in order to achieve different tuning
step sizes in different frequency bands are described in U.S. Pat.
Nos. 5,535,432 and 5,610,559 which are also hereby incorporated
by reference.
Another approach is to employ an offset voltage controlled oscillator
(VCO) to mix up or down the receiver local oscillator frequency
to generate the transmitter frequency. This approach is disclosed,
for example, in commonly assigned U.S. patent application Ser. Nos.
08/675,171 and 08/823,068.
U.S. patent application Ser. No. 08/675,171, entitled "Dual
Band Transceiver", describes the use of common radio components
for dual bands. Sharing a common receiver local oscillator synthesizer,
transmitter offset oscillator loop, transmitter UHF VCO, IF filter,
and receiver IF circuit greatly reduces the number of components
for a dual band mobile phone. However, the transmitter offset frequency
is fixed for both bands. Thus, one band has to jog the main channel
stepper to put the transmitter on frequency, i.e., one band has
to change frequency for transmission. The band that must do the
hop cannot, therefore, be used in full duplex mode, which is required
for CDMA and multi-rate TDMA mobile phones.
U.S. patent application Ser. No. 08/823,068, entitled "Dual
Band Mobile Station", also describes the sharing of similar
common radio components between bands but enables duplex operation
in both bands. However, as with all offset schemes, generation of
the transmitter carrier is inherently troublesome because of the
spurious performance of the transmitter. This is because mixing
the synthesized VHF signal up with the receiver local oscillator
frequency to the desired transmit band spawns up-conversion products
which have to be filtered. These products may add a filtering burden
to meet the transmitter output spectral mask requirements.
Continuous advances in electronics allow for smaller mobile phones
complying with a variety of national and international protocols.
The international mobile phone standard known as GSM in Europe and
as PCS 1900 in the USA operates with a transmit/receive duplex spacing
of 45 MHz in the European 900 MHz band; 95 MHz in the European 1800
MHz band, and 80 MHz in the U.S. 1900 MHz PCS band. The channel
spacing is 200 KHz (13 MHz/65) and the transmitted symbol rate is
13 MHz/48. All timing in this standard is related to a 13 MHz clock,
as is well known. The U.S. IS 136 system known as DAMPS operates
with a 45 MHz duplex spacing in the US 800 MHz cellular band, and
with an 80.4 MHz duplex spacing in the U.S. 1900 MHz PCS band, with
a tuning step size of 30 KHz and a transmitted symbol rate of 24.3
Kilosymbols/sec. In IS 136, as is well known, the tuning step sizes
and symbol rates and internal timing are all derivable from a 19.44
MHz clock. Yet another U.S. standard known as IS95 uses Code Division
Multiple Access at a transmitted chip rate of 1228.8 MHz, with a
duplex spacing of 45 MHz combined with tuning steps of 30 KHz in
the 800 MHz band, alternatively 50 KHz steps combined with 80 MHz
duplex spacing in the 1900 MHz band. In IS95, the chip rate and
frequency step sizes are not easily derivable from the same crystal
oscillator.
U.S. Pat. No. 5,471,652 discloses an arrangement including a multiplier
which permits the operating point of a VCO in a single band phone
to be adjusted for various bands. This patent does not discuss application
to a dual band phone. In addition, according to this patent, a synthesized
VHF signal (80-400 MHz) must be up-converted to the transmitter
channel for digital (non-FM) operation. In the FM mode, the UHF
oscillator is directly modulated and bled through the mixer.
It may be easily understood that combining two or more of the various
abovementioned protocols in the same handheld unit is hindered by
the variety of tuning step sizes, duplex spacings and symbol rates
that must be synthesized. Consequently, there exists a need for
an improved radio architecture to facilitate such combination.
SUMMARY OF THE INVENTION
It is therefore an object of the present invention to provide a
transceiver which supports full duplex operation in dual bands.
It is a further object of the present invention to minimize the
generation of spurs, thereby reducing the filtering design requirements
of the transmitter. It is still a further objection of the present
invention to provide a dual band transceiver that can accommodate
multiple standards within the same handset.
According to a first embodiment of the invention, a mobile phone
receiver comprises a first superheterodyne down conversion means
using a first local oscillator frequency which can be tuned in frequency
steps by a programmable digital frequency synthesizer phase lock
loop (PLL). The first down conversion means converts received signals
to a first intermediate frequency (IF) for filtering. A second down
conversion means using a second local oscillator converts first
IF signals to a second IF or to the complex baseband for further
filtering and processing. The second local oscillator is generated
using a second digital frequency synthesizer PLL which locks the
second oscillator to a crystal reference oscillator. The crystal
reference oscillator provides a buffered clock output signal from
which digital logic derives transmit symbol rates and receiver processing
sampling rates.
According to a first aspect of the invention, the second local
oscillator provides a buffered output signal at the second local
oscillator frequency. The buffered output signal is used as the
reference frequency for the first local oscillator's synthesizer
PLL, thus eliminating the need to distribute the crystal oscillator
signal to the first oscillator's PLL circuit. According to a second
aspect of the invention, the first oscillator PLL comprises a phase
comparator to compare the divided down first local oscillator signal
with the divided down reference frequency signal from the second
local oscillator, the divided down frequencies being equal to the
desired receiver frequency tuning steps or a multiple thereof. It
should be appreciated that this frequency would not have been available
by dividing down the crystal frequency in an integral ratio without
practicing this aspect of the invention.
According to a third aspect of the invention, a third digital frequency
synthesizer PLL controls the transmitter frequency to be equal to
the first local oscillator frequency plus or minus a transmit offset
frequency. The transmit frequency can for example be heterodyned
with the first local oscillator frequency to produce a transmit
offset frequency signal; the transmit offset frequency signal is
then divided down in a digital divider and compared with a phase
reference frequency which is also derived by dividing the second
local oscillator frequency by an integer factor.
Since according to the third aspect of the invention, the transmit
offset synthesizer PLL and the first local oscillator PLL both utilize
the second local oscillator as a common frequency reference, they
can furthermore be packaged in a common integrated circuit and can
share at least part of the reference divider which divides the second
local oscillator frequency to produce a first and second phase comparator
reference frequency signal for the two PLLs respectively. The two
PLLs' respective phase comparators are arranged to respond to opposite
polarities of a signal at a lowest common multiple frequency of
their respective first and second phase comparator reference signals
in order to minimize mutual interference between the two PLLs.
According to a second embodiment of the invention, a mobile phone
receiver comprises a first down conversion means using a first local
oscillator frequency which can be tuned in frequency steps by a
digital frequency synthesizer PLL which is locked to a reference
frequency. The first down conversion means converts received signals
to a first IF for filtering. A second down conversion means using
a second local oscillator converts first IF signals to a second
IF. The second local oscillator frequency is generated using a second
digital frequency synthesizer PLL which locks the second oscillator
to the reference frequency. A third down conversion means mixes
the transmit frequency with the first local oscillator frequency
to produce a lock frequency. A third digital frequency synthesizer
PLL compares the lock frequency and the reference frequency to control
generation of the transmit frequency.
BRIEF DESCRIPTION OF THE DRAWINGS
The invention will be more fully understood upon reading the following
description with reference to the accompanying drawings, in which:
FIG. 1 illustrates prior art reference frequency distribution;
FIG. 2 illustrates another prior art scheme;
FIG. 3 illustrates the improved system described in the incorporated
Dolman reference;
FIG. 4 illustrates a reference distribution scheme according to
a first embodiment;
FIG. 5 illustrates more detail of the inventive frequency synthesis
scheme according to the first embodiment;
FIG. 6 illustrates an inventive dual band scheme using frequency
doublers according to the first embodiment;
FIG. 7 illustrates an inventive dual band scheme using frequency
halvers according to the first embodiment;
FIG. 8 illustrates an inventive scheme using one frequency halver
and one frequency doubler according to the first embodiment;
FIG. 9 illustrates an inventive scheme in which the frequency halver
and the frequency doubler are switched in position according to
the first embodiment;
FIG. 10 illustrates an inventive scheme in which a frequency doubler
can be powered down for AMPS reception according to the first embodiment;
FIG. 11 is a modification of FIG. 10, using frequency halvers;
FIG. 12 illustrates an alternative reference frequency distribution
for dual-mode radios according to the first embodiment;
FIG. 13 illustrates divider ratios for the dual-mode radio of FIG.
12;
FIG. 14 illustrates a dual-mode radio using I.F. Homodyne for a
PCS1900 mode according to the first embodiment;
FIG. 15 illustrates a dual-mode radio using a single crystal according
to the first embodiment;
FIG. 16 illustrates divider ratios for the dual-mode radio of FIG.
15;
FIG. 17 illustrates a dual-mode radio using two reference crystals
according to the first embodiment;
FIG. 18 illustrates divider ratios for the dual-mode radio of FIG.
17;
FIG. 19 illustrates divider ratios to eliminate the second crystal
of FIG. 17;
FIG. 20 illustrates a skip-counter for generating 194.4 KHz from
19.5 MHz according to the first embodiment;
FIG. 21 illustrates a dual band scheme according to a second embodiment
of the present invention;
FIG. 22 illustrates more detail of the inventive dual band scheme
using a filter according to the second embodiment;
FIG. 23 illustrates an inventive dual band scheme using narrow
band modulators according to the second embodiment;
FIG. 24 illustrates an inventive dual band scheme using a frequency
divider according to the second embodiment;
FIG. 25 illustrates an inventive dual band scheme using various
frequency multipliers/dividers according to the second embodiment;
and
FIG. 26 illustrates an inventive dual band scheme using an alternative
arrangement of variable gain amplifiers according to the second
embodiment .
DETAILED DESCRIPTION
Referring now to FIG. 1, a prior art cellular phone comprises an
antenna (10) connected to a receiver and a transmitter by means
of a transmit/receive duplexor (11). When simultaneous transmit
and receive (frequency duplex) is used, as in the analog FM AMPS
standard or the IS95 CDMA standard, the duplexor (11) is a duplexing
filter. Alternatively, for a TDMA system such as GSM/PCS1900 or
D-AMPS/IS136 that employs time-duplex, the duplexor can be a T/R
switch. For dual band phones employing frequency duplex in one band
and time duplex in another band, duplexor (11) can be a dual-band
duplexor having both a switch and a duplexing filter. When frequency
duplex is used in both bands, duplexor (11) could comprise duplexing
filters for both bands, and when time duplex is used in both bands,
a single T/R switch might serve for both bands.
The duplexor allows the transmitter to be connected to the antenna
without affecting receiver sensitivity. The receiver comprises a
low noise amplifier and downconvertor known as "the front end",
(12). The front end can be fabricated in a single integrated circuit
comprising a low-noise amplifier, a downconverting and possibly
image-rejecting mixer and a first local oscillator, for each of
two or more different frequency bands (such as the 800 MHz and 1900
MHz bands).
The first local oscillator mixes with the desired receive frequency
signal to produce a first intermediate frequency signal. Filtering
can take place with a fixed frequency bandpass filter, IF filter
(15). The desired receive frequency is selected by tuning the local
oscillator to a frequency equal to the sum or the difference of
the desired receive frequency and the first IF, by means of first
local oscillator synthesizer phase lock loop (14). The 1st LO PLL
tunes the 1st LO to a programmable integer multiple of a basic tuning
step size, which is derived from a crystal reference oscillator
(21) by dividing the crystal frequency by another integer to get
the step size. For small step sizes, the synthesizer may alternatively
derive a greater step size from crystal oscillator (21) by dividing
by a smaller integer, and then interpolate between these larger
steps to get the desired smaller steps using the technique of fractional-N
synthesis described in the above-incorporated references. First
LO PLL circuit (14) compares the 1st LO frequency to the crystal
reference signal and generates an error signal. The error signal
is filtered and integrated in Loop Filter (24) to produce a control
signal to control the oscillator frequency until the frequency is
exactly as intended.
The receiver amplifies the filtered 1st IF signal and then customarily
performs a second frequency down conversion using a second heterodyne
mixer and second local oscillator. The IF amplifiers, second local
oscillator and second mixer are all contained in a conventional
second integrated circuit (16). After downconverting a second time
to a second or final Intermediate frequency, further amplification
may take place at the final IF and a detector circuit can be employed
to produce a Radio Signal Strength Indication (RSSI) related to
the strength of the received signal. The second IF amplifier may
be hardlimiting, and then outputs a hardlimited final IF signal
to digital signal processing (20) where phase information is extracted
and digitized using the second IF signal simultaneously with digitizing
the RSSI signal, as described in U.S. Pat. No. 5,048,059 entitled
"Log-polar signal processing", which is hereby incorporated
by reference herein. The second local oscillator part of IF amplifier
circuit (16) is also controlled to the desired frequency by means
of a synthesizer PLL circuit (17) and loop filter (23). The 2nd
LO frequency is compared with the crystal oscillator (21) and an
error signal produced as before. Thus both synthesizer circuits
(14) and (17) use the crystal as the frequency reference or standard
of accuracy for controlling both the first and second LO. The digital
signal processing logic (20) can also require an accurate frequency
standard for producing receiver sampling and processing rates and
transmit symbol rates, and so is also fed with an output from crystal
oscillator (21).
The transmitter comprises transmit frequency generation circuit
(19) for producing a signal offset from the receive frequency by
the fixed duplex spacing. The transmit frequency is thus offset
from the 1st LO frequency by the duplex spacing combined with the
first intermediate frequency, which nevertheless is still a constant
offset. The constant transmit offset is either equal to the first
IF minus the duplex spacing or the first IF plus the duplex spacing,
depending on whether the 1st LO is lower or higher than the receive
and transmit frequencies.
The transmit frequency signal is then modulated with information
from digital signal processor (20) using modulator (18), which is
for example a quadrature modulator having I and Q input signals.
The modulated signal is then amplified to a transmit power level
using power amplifier (13), which may be a dual-band power amplifier
in a dual-band phone.
Transmit offset PLL forms the difference between the transmit and
1st LO frequencies and tests to see if it equals the desired offset,
by comparing the offset with the crystal reference. The TX offset
PLL thus also has need of a crystal reference frequency signal from
oscillator (21), making four places to which the oscillator signals
must be distributed.
The four separate outputs from oscillator (21) must be sufficiently
isolated from one another by buffer amplifiers and conditioned to
drive copper tracks on the printed circuit motherboard. This consumes
battery power and presents a radiated interference hazard. Often,
to save battery power during standby, outputs which are momentarily
not required, such as that feeding the transmit offset PLL, may
be turned off by control signals from a control processor (part
of digital signal processing 20), which is a further complication.
It is thus desirable to reduce the distribution of the crystal reference
signal by means of printed circuit board tracks to multiple destinations.
A first step in this direction is already taken in prior art products
sold by Ericsson in the USA. By combining the first and second LO
PLLs in a single chip, such as the Philips UM1005 or 8026 part,
a single input for the crystal reference may be used, since both
use the crystal as the reference. Further, by combining crystal
oscillator (21) into a transmit signal generator chip together with
TX offset PLL and modulator (18), no external output connection
between oscillator (21) and offset PLL (19) is needed.
Thus, referring to FIG. 2, the number of crystal reference signal
outputs required is reduced to two, one feeding dual synthesizer
circuit (14+17) and the other feeding digital signal processing
(20).
In U.S. patent application Ser. No. 08/795,930 to Dolman, incorporated
above, it is explained that all PLLs desirably operate by dividing
the crystal reference frequency by the smallest possible integer
for comparison with the frequency of the oscillator they are controlling,
which is also divided by the smallest possible integer. Expressing
this another way, it is desired to have the largest possible common
factors between the controlled oscillator frequencies and the reference
frequency. Dolman discloses that this is facilitated when the transmit
offset frequency is generated using the 2nd LO as the reference
frequency rather than the crystal (22). Dolman's inventive arrangement
is shown in FIG. 3.
The second LO provides a first output signal to its controlling
PLL (17) and a second output to TX offset PLL (19). Since the crystal
oscillator is now not used for any purpose in transmit circuits
(18,19) oscillator (21) is shown once more as a separate circuit
(21), having two buffered outputs. The total number of radio frequency
signal outputs has however increased, as has the number of separate
integrated circuit chips. The radio frequency signals distributed
on the printed circuit board are as follows:
1) 1st LO signal from frontend (12) to PLL (14);
2) 1st LO signal from frontend (12) to TX offset PLL (19);
3) Crystal reference frequency from oscillator (21) to PLL (14+17);
4) Crystal reference frequency from oscillator (21) to processing
(20);
5) 2nd LO from IF chip (16) to controlling PLL (17); and
6) 2nd LO from IF chip (16) to TX offset PLL (19).
It is an object of the current invention to reduce the number of
RF distribution tracks from the six enumerated above.
FIG. 4 shows one implementation of the invention. The two places
to which the 1st LO signal is routed, that is the first LO PLL (14)
and the TX offset PLL (19) are colocated with modulator circuit
(18) in first (transmit) integrated circuit. Thus there is only
a single 1st LO output connection from front end chip (12) to transmit
chip (14,18,19).
However, when two synthesizer PLLs are colocated on the same chip,
they should produce output pulses at different times. This is difficult
or impossible to arrange if the two phase comparators have independent
reference frequency sources, such as when the second LO is used
for the TX offset PLL reference and the crystal is used as the first
LO reference. Therefore, according to this invention the second
LO is also used as the reference for the first LO synthesizer PLL.
Moreover, as will be shown below, there are numerical advantages
in using the second LO as the reference source for the first LO,
particularly when it is desired to construct dual-band/dual-mode
radios. Thus, a single reference input from 2nd LO part of IF chip
(16) is provided for both PLLs (14) and (19).
Crystal oscillator circuit (21) is now combined with 2nd LO PLL
and IF circuit (16) so that the reference signal from oscillator
(21) to PLL (17) is an internal connection only. Likewise, the 2nd
LO signal to its controlling PLL (17) is an internal signal only.
The only remaining external signal is from reference oscillator
(21) to digital processing (20).
The radio frequency oscillator signal distribution has now been
reduced to the following signals:
1) 1st LO signal from frontend (12) to PLLs (14 & 19);
2) Crystal reference frequency from oscillator (21) to processing
(20); and
3) 2nd LO from IF chip (16) to TX offset PLLs (14 & 19).
It would have been equally conceivable to place crystal oscillator
(21) into digital processing chip (20), however, the oscillator
(21) is more logically associated with other analog/RF circuits
that use the same integrated circuit fabrication processes, and
is therefore envisioned to be preferably integrated with IF chip
(16,17,21). It is possible that in some applications a Very High
Frequency (VHF) crystal, such as an overtone crystal, could be used
directly to control the frequency of the second local oscillator
without the use of a digital frequency synthesizer PLL circuit;
however, VHF overtone crystals are more difficult to adjust to a
desired oscillation frequency than fundamental-mode crystals, so
a fundamental-mode crystal reference oscillator with a digital PLL
is preferred.
FIG. 5 gives more details of the reference frequency distribution
and frequency synthesis arrangements according to the inventive
block diagram of FIG. 4.
The basic source of an accurate frequency reference in the apparatus
is quartz crystal resonator (22) of FIGS. 1-4, connected to oscillator
circuit (21). Even a quartz crystal cannot provide the necessary
accuracy required for cellular phones operating in the 2 GHz region
of the radio spectrum, and therefore means contained within digital
processing (20) determine the receiver frequency error relative
to signals received from land-based network stations or satellite
relays, which error is ascribed to crystal (22), and an adjusting
signal is sent to frequency adjustment components (such as a varactor
diode for example) connected to crystal (22) so as to annul the
error.
In FIG. 5, the oscillator circuit (21) is incorporated in IF chip
(30) along with second local oscillator (33) and its control PLL
comprising reference divider (35), first variable divider (32),
phase comparator (31) and loop filter (34). The crystal oscillator
signal is divided in frequency by counter/divider (35) which divides
by a first integer M1 to produce a phase comparison frequency F.sub.ref
/M1, where F.sub.ref is the crystal frequency. The second local
oscillator signal is divided in frequency by an integer N1 in first
variable divider (32), to produce a second phase comparison signal,
which is compared with the phase comparison frequency signal from
divide-by-M1 circuit (35) to produce a phase and frequency error
signal from first phase comparator (31). The phase error signal
is filtered and integrated using loop filter (34) to produce a frequency
control signal to 2nd local oscillator (33) free of comparison frequency
ripple. The higher the comparison frequency from divider (35), the
easier it is for loop filter (34) to eliminate this unwanted ripple
while otherwise maintaining a fast speed of response to correct
unwanted fluctuations of the 2nd LO's frequency due to noise or
vibration, for example. Therefore an objective of the invention
is to obtain a high comparison frequency, that is a low reference
divide ratio M1. The second local oscillator frequency is thus accurately
controlled to equal F.sub.ref.N1/M1.
According to Dolman's prior invention and the current invention,
a buffered second local oscillator signal is output from 2nd LO
(33) to be used as a reference for other frequency generation, in
particular the transmit offset frequency (according to Dolman's
prior application referenced above) and now also the first local
oscillator frequency according to this invention. Since both the
TX offset and 1st LO synthesizer PLL circuits are to be colocated
in transmit signal generation chip (40) in order to reduce distribution
of the second LO signal to a single cross-board connection, it is
desirable that the phase comparators of the respective PLLs should
pulse at different times separated as far as possible within some
longest common period. This ensures that when one charge-pump phase
comparator receives a current pulse from the supply, the other charge
pump is in its tristate, that is, a high impedance state or open
circuit output, with no current flowing to its respective loop filter.
This minimizes the risk of interference from one charge pump to
the other. The design and operation of charge pump phase detectors
is described more fully in above-incorporated U.S. Pat. No. 5,095,288.
In order to provide the preferred out-of-phase relationship between
the charge pumps (43,49), internal frequency plans are sought in
which the phase comparison frequency for the TX offset loop is an
integer multiple M3 of the phase comparison frequency for the first
LO. Since the second LO frequency is also, according to Dolman's
incorporated application, an integer M2 times the TX offset reference,
the first LO comparison frequency must now be related to the second
LO frequency divided by M2.M3.
Thus the second LO frequency signal is input from IF chip (30)
to TX chip (40) and divided by integer M2 in a second reference
divider (41) to obtain the phase reference to TX offset phase comparator
(43) according to Dolman as
This frequency is then further divided in third reference divider
(42) by integer M3 in order to obtain the phase comparison frequency
for first LO phase comparator (49). Moreover, divider M3 and phase
comparator (43) are arranged to respond to opposite edges of the
output of divider M2, for example one responding to a rising edge
(low voltage or `0` state transitioning to high voltage or `1` state)
while the other responds to a falling edge (1 to 0 transition).
This ensures that they respond one half cycle apart in time of their
lowest common multiple frequency at the output of divider (41).
The phase comparison rate for charge pump phase comparator (49)
is thus
This frequency is compared with the 1st LO frequency from first
LO (51) divided down by a factor N3 in third variable divider (48),
to produce a frequency and phase error signal from comparator (49)
which is filtered in loop filter (52) to obtain a feedback control
signal to control oscillator (51) to the desired 1st LO frequency
Preferably, N3 is not an integer factor but comprises a whole part
and a fractional part, the components (48,49 and 52) of the first
LO PLL forming a fractional-N synthesizer according to above-incorporated
Pat. No. 5,180,993. Optionally, both M3 and N3 can be varied in
a pattern generated by a fractional-(N,M) controller according to
U.S. patent application Ser. No. 08/957,173 entitled "Frequency
Synthesis by Sequential Fraction Approximations" (Dent, filed
Oct. 24, 1997) which is hereby incorporated by reference. Both the
fractional-N and the fractional-(N,M) techniques have the desirable
effect of allowing the 1st LO phase comparison frequency to be higher
than the desired tuning step size, thus making it easier for the
loop filter (52) to filter out unwanted comparison frequency ripple
while otherwise maintaining a fast control loop response to correct
errors.
The transmitter frequency signal, when transmission is required,
is generated by transmit frequency oscillator (45). A transmit frequency
signal from oscillator (45) is mixed in TX mixer (46) with a first
LO signal from first LO (51). The first LO signal preferably comes
from receive chip (12) via a single cross board connection to minimize
RF tracks. A single cross board connection for any of the interchip
signals mentioned can nevertheless be a balanced connection comprising
two tracks driven in antiphase, as balanced connections to and from
RF chips at high frequencies reduce unwanted stray coupling and
radiation effects.
TX mixer (46) mixes the transmit frequency and the 1st LO frequency
to produce a difference frequency signal at the TX offset frequency
F.sub.txoff. The difference frequency signal output from mixer (46)
may be low-pass filtered to ensure the original, higher input frequencies
are removed, and then drives second variable divider (47) which
divides by a factor N2. The output signal at frequency F.sub.txoff
/N2 is then compared in second phase comparator (43) with the phase
reference from divider (41) to produce a frequency and phase error
signal. The error signal from comparator (43) is filtered and integrated
in loop filter (44) to produce a control signal which controls TX
oscillator (45) until the desired TX-offset frequency is accurately
achieved. Thus the TX offset frequency is given by
It is possible for the TX offset PLL comprising components (41,43,44,45,46,47)
also to be a fractional-N synthesizer, however fractional-N synthesizers
are more complex than integer synthesizers and thus it is desired
to avoid having more than one in the apparatus. Thus, factor N2
is preferably an integer.
Obtaining the highest possible phase comparison frequencies for
phase comparators (31,43,49) is rarely a problem in single-band
radios having a single duplex spacing between transmit and receive
frequency channels. It is more difficult first in the context of
two-band radios that must operate with more than one duplex spacing.
Therefore two-band radio designs according to the invention will
now be described with the aid of FIGS. 6,7,8 and 9.
A two-band radio according to FIG. 6 would comprise a transmit
frequency oscillator (45) for generating transmit frequencies in
the lower of the two possible transmit frequency bands, A frequency
doubler (45a) is then used to double the frequency when operation
in the higher of the two frequency bands is desired, the lower and
higher bands being approximately one octave apart. An output from
the oscillator (45) directly is used to drive the modulator when
lower-band operation is required while an output from the doubler
(45a) is used when operation at the higher band is desired. However,
as indicated in FIG. 6, the lower frequency direct from oscillator
(45) enters TX mixer (46).
Likewise, first local oscillator (51) operates on a frequency adapted
to the lower of the two-possible receive frequency bands to convert
received signals to the desired first Intermediate Frequency; the
signal from first LO (51) is doubled in frequency using doubler
(51a) when operation in the higher of the two receive frequency
bands is desired, the LO frequency for the higher band being approximately
an octave higher than that for the lower band. This approximation
can be manipulated to be a close approximation by suitable choice
of the first Intermediate frequency and by suitable choice of either
high-side or low-side mixing in front-end chip (12).
For example, for low-band receive operation, we have ##EQU1## F.sub.101
(lo) is the low-band first LO frequency, F.sub.rx (lo) is the low-band
receive channel frequency and
F.sub.if1 is the chosen first intermediate frequency. ##EQU2##
F.sub.101 (hi) is the high-band first LO frequency, F.sub.rx (hi)
is the high-band receive channel frequency and
F.sub.if1 is the same chosen first intermediate frequency as for
low-band.
Thus, for F.sub.101 (hi) to be twice F.sub.101 (lo), we have ##EQU3##
The latter two equations give negative results which is impossible.
A possible alternative is to make the first Lo range at high band
triple the first LO range at low band, giving ##EQU4## Examples
of preferred internal frequency plans for a radio according to FIGS.
4 and 5, operating according to the IS54 "D-AMPS" single-band
standard, will now be described. A search of frequency plans giving
the highest possible phase comparison frequencies at phase comparators
(31,43 and 49) yielded, among others, the results:
__________________________________________________________________________
1.sup.ST IF 2.sup.ND LO TXOFFSET M1 N1 M2 N2 M3 1.sup.ST LO FRAC-N
MODULUS __________________________________________________________________________
101.64 101.52 146.64 9 47 9 13 47 8 __________________________________________________________________________
The above results provide phase comparison frequencies of
F.sub.xtal /M1=19.44/9=2.16 MHz for 2nd LO phase comparator (31);
F.sub.lo2 /M2=101.52/9=11.28 MHz for TX offset phase comparator
(43), and
F.sub.lo2 /(M2.M3)=11.28/47=240 KHz for 1st LO phase comparator
(49).
The first LO tuning steps are reduced from the above 240 KHz to
30 KHz by employing a fractional-N divider for N3 giving steps of
1/8th, i.e. the fractional-N modulus is 8.
The above solution provides a high TX offset phase comparison frequency
of 11.28 MHz. Other criteria might be to obtain the highest 2nd
LO phase comparison frequency. An alternative result for which the
2nd LO is just a harmonic of the crystal is for example:
__________________________________________________________________________
1.sup.ST IF 2.sup.ND LO TXOFFSET M1 N1 M2 N2 M3 1.sup.ST LO FRAC-N
MODULUS __________________________________________________________________________
116.76 116.64 161.76 1 6 243 337 1 16 __________________________________________________________________________
The above values result in a 2nd LO phase comparison frequency
of 19.44 MHz at phase comparator (33), and divider (35) is not necessary
because M1=1. The transmit offset and first LO phase comparators
(43,49) both operate at 480 KHz, and divider (42) may be omitted
as M3=1. The first LO tuning steps are reduced from 480 KHz to 30
KHz by employing a modulus 16 fractional-N divider (48) allowing
N3 to be varied in steps of 1/16th.
Attention is now turned to dual-band radios with internal frequency
reference distribution according to FIG. 6. The above two exemplary
solutions were illustrated because they are also compatible with
a dual-band radio operating according to the dual-band D-AMPS standard
IS136. Solutions for dual-band radios are given in the tables below
for cases where the first local oscillator is on the high side for
800 MHz hand operation and on the low side for 1900 MHz band operation,
and the second IF is fixed at 120 KHz.
Table 1 illustrates solutions in which the 2nd LO is a harmonic
of the crystal, that is the second LO has the highest possible phase
comparison frequency, M1 being equal to unity.
TABLE 1 __________________________________________________________________________
Dual band 800 (1900) solutions with 2nd LO a crystal harmonic 1st
IF 2nd LO TX OFFSET M1 N1 M2 N2 M3 1st LO modulus __________________________________________________________________________
116.76 116.64 161.76 (36.72) 1 6 243 (54) 337 (17 ) 1 (9 or 3) 16
(8 or 24) 155.64 155.52 200.64 (75.6) 1 8 162 (72) 209 (35) 1 (9
or 3) 32 (8 or 24) 194.52 194.4 239.52 (114.48) 1 10 405 (90) 499
(53) 1 (9 or 3) 16 (8 or 24) 233.4 1233.28 278.4 (153.36) 1 12 243
(108) 290 (71) 1 (9 or 3) 32 (8 or 24) __________________________________________________________________________
When using the figures in table 1 above to determine the phase
detector comparison frequencies for TX offset phase comparator (43),
it must be taken into account that the arrangement in FIG. 6 controls
the TX oscillator (S1) frequency BEFORE doubling to 1900 MHz in
doubler (51a).
Therefore, phase comparator (43) must operate at half the frequency
provided by divider (41) and the indicated value of M2.
Thus, either phase comparator (43) must contain a further divide
by two circuit to halve the frequency from divider (41) when operating
in the 1900 MHz band, or else the value of M2 for 1900 MHz must
be doubled.
In the latter case, the value of M3 for 1900 MHz operation must
be halved (which is impossible as M3 is always odd at 1900 MHz),
or else the fractional modulus for 1900 MHz operation must be halved.
The latter is preferable and so the preferred fractional-modulus
at 1900 MHZ is 4 or 12 combined with an M2 value double that shown
in table 1 for 1900 MHz operation. The TX offset phase comparison
frequency at phase comparator (43) is thus 1080 KHz for 1900 MHz
operation and not 2160 KHz as would be obtained by dividing the
second LO frequency in table 1 by the indicated values of M2.
Furthermore, note that in FIG. 6 it is always the doubled frequency
from doubler (S1a) that is fed to fractional-N first-LO synthesizer
loop beginning with variable divider (48) for N3. Since the frequency
used for the receiver mixer during 800 MHz band operation is half
of the synthesized frequency, the synthesizer need only provide
60 KHz steps in order to tune the receiver in 30 KHz steps. Thus
the fractional-N modulus shown in table 1 for 800 MHz operation
may be halved.
It may be desirable to operate in both bands using the same fractional-N
modulus, and this may always be accomplished by use of a modulus
which is the lowest common multiple of the 800 MHz and 1900 MHz
moduli, accepting that the frequency steps in one or both bands
may then be finer than needed, it being acceptable to exceed the
required frequency resolution.
The above issues are one motivation for considering frequency halving
circuits (45b,S1b) in FIG. 7 as opposed to frequency doubling circuits
(45a,51a) of FIG. 6. Another motivation is that phase noise is doubled
by frequency doubling circuits but halved by frequency halving circuits.
Thus there is an expectation of lower unwanted phase noise and ripple
when using frequency halving circuits. Yet another motivation is
that a frequency doubling circuit requires a filter to remove unwanted
leakage of the fundamental, as well as other unwanted higher harmonics;
the output of a frequency divide-by-2 circuit is however relatively
free of other unwanted spectral components.
Referring now to FIG. 7, it is seen that it is the non-divided
output of oscillator (S1) which is fed to the TX offset synthesizer
loop beginning with mixer (46). Therefore the phase comparator for
800 MHz band operation must operate at double the frequency implied
by table 1, i.e. the value of M2 must be halved, alternatively the
value of N2 must be double that shown in table 1 for 800 MHz operation.
The former is not possible for cases where M2 is odd, but is possible
when the first IF is 155.64 MHz and M2=162. Thus, when table 1 is
applied to FIG. 7, the values of N2 for 800 MHz should be doubled
except in the case of first IF=155.64 MHz, in which case a better
option is to halve M2 to 81; then it is necessary to double the
value of M3 (to two) for 800 MHz operation in order to maintain
the same 1st LO phase comparison frequency at phase comparator (49),
alternatively to increase the fractional-N modulus from 32 to 64.
On the other hand, since the frequency of oscillator (51) is halved
before use in the receiver for 800 MHz operation, it is sufficient
that oscillator (51) be tuned in 60 KHz steps, allowing the fractional-N
modulus to be halved again back to 32.
The above considerations with respect to FIG. 7 also apply to the
arrangement of FIGS. 8 and 9, in which the TX frequency signal and
the first LO are always controlled at the higher frequency and halved
for 800 MHz use.
In choosing between the implementations of FIGS. 6,7,8 and 9, another
motivation is power consumption. In FIG. 6, the x2 circuit 51a must
be powered up during 800 MHz band receive operation, which has most
impact on standby time before the battery must be replenished. Still,
the x2 circuit 45a need only be powered up for 1900 MHz transmission,
thus saving power in 800 MHz band transmission. In FIG. 7, divider
51b need only be powered up for 800 MHz reception, and may be powered
down for 1900 MHz reception. Divide by 2 circuit 45b likewise need
only be powered up for 800 MHz transmission and is not needed for
1900 MHz transmission.
In FIG. 8, x2 circuit 45a must be powered up for transmission in
either frequency band, but this is of little consequence as the
power amplifier (13) dominates transmit power consumption. Divide
by 2 circuit 51b may be powered down during 1900 MHz receive. In
FIGS. 6 and 9, x2 circuit 51a must always be powered up for reception
in either frequency band. Therefore this is not as desirable for
standby battery life at 1900 MHz as FIGS. 7 or 8.
1900 MHz D-AMPS operation uses TDMA, which affords much longer
standby times due to a low receive duty factor. 800 MHz operation
however includes the analog FM AMPS mode, in which receive standby
duty factor is longer. The 800 MHz AMPS operation is therefore the
limiting factor for battery life and we are therefore led to consider
FIG. 10, in which the first LO is always controlled at the lower
frequency, allowing doubler 51A to be powered down during 800 MHz
reception.
Referring to FIG. 10, the first LO is always controlled at the
lower frequency, i.e. before doubling. This allows doubler 51a to
be powered down in 800 MHz operation. A disadvantage, however, is
that the oscillator 51 must be tuned in 15 KHz steps in order to
provide 30 KHz steps at 1900 MHz, requiring the fractional-N modulus
to be doubled, which is undesirable. Taking into account that a
divide-by-2 circuit in present semiconductor technology consumes
very little power, and likely less than a frequency doubler circuit,
together with the other advantages outlined above for frequency
halving rather than frequency doubling, FIG. 7 is likely to be the
best practical implementation.
The above Table 1 listed solutions for which the second LO was
a harmonic of the crystal, giving the lowest value of unity for
divider (35). Table 2 lists solutions in which 2nd LO phase comparator
(31) operates at 6.48 MHz, which is the crystal frequency divided
by 3 (M1=3).
TABLE 2 __________________________________________________________________________
Dual band 800 (1900) with 2nd LO a multiple of crystal/3 1st IF
2nd LO TX OFFSET M1 N1 M2 N2 M3 1st LO modulus __________________________________________________________________________
90.84 90.72 135.84 (10.8) 3 14 189 (42) 283 (5) 1 (9 or 3) 16 (8
or 24) 103.8 103.68 148.8 (23.76) 3 16 108 (48) 155 (11) 1 (9 or
3) 32 (7 or 24) 129.72 129.6 174.72 (49.68) 3 20 1135 (60) 182 (23)
1 (9 or 3) 32 (8 or 24) 142.68 142.56 187.68 (62.64) 3 22 297 (66)
391 (29) 1 (9 or 3) 16 (8 or 24) 168.6 168.48 213.6 (88.56) 3 26
351 (78) 445 (41) 1 (9 or 3) 16 (8 or 24) 1181.56 181.44 226.56
(101.52) 3 28 189 (84) 236 (47) 1 (9 or 3) 32 (8 or 24) 207.48 207.36
252.48 (127.44) 3 32 216 (96) 263 (59) 1 (9 or 3) 32 (8 or 24) 220.44
220.32 265.44 (140.4) 3 34 459 (102) 553 (65) 1 (9 or 3) 16 (8 or
24) __________________________________________________________________________
There are also many solutions with the 2nd local oscillator a multiple
of 2.16 MHz (crystal/9, i.e. M1=9), or 720 KHz (crystal/27 or M1=27),
and at least one solution with M1=6. Table 3 below only lists other
solutions that have particularly interesting characteristics such
as high comparison frequencies for TX offset comparator (43) in
either 800 MHz or 1900 MHz operation.
TABLE 3 __________________________________________________________________________
Other solutions of particular interest 1st IF 2nd LO TX OFFSET M1
N1 M2 N2 M3 1st LO modulus __________________________________________________________________________
152.4 152.28 197.4 (72.36) 6 47 27 (141) 35 (67) 47 (3) 4 (12) 101.64
101.52 146.64 (21.6) 9 47 9 (47) 13 (10) 47 (9 or 3) 8 (8 or 24)
159.96 159.84 204.96 (79.92) 9 74 333 (2) 427 (1) 1 (333 or 111)
16 (8 or 24) 203.16 203.04 248.16 (123.12) 9 94 9 (94) 11 (57) 47
(9 or 3) 16 (8 or 24) 106.68 106.56 151.68 (26.64) 27 148 111 (4)
158 (1) 1 (111 or 37) 32 (8 or 24) 213.24 213.12 258.24 (133.2)
27 296 222 (8) 269 (5) 1 (111 or 37) 32 (8 or 24) __________________________________________________________________________
The above solutions are remarkable for their relatively low values
of (M2,N2) in one or other frequency band, giving very high TX offset
phase comparison frequencies in those cases.
As described above, it is of interest that the first local oscillator
in the higher frequency range should tune over a range approximately
equal to twice the range of frequencies needed for operation in
the lower band.
The receive frequency range for the 800 MHz cellular band is 869.04
to 893.97 MHz, while the receive frequency range of the 1900 MHz
PCS bands is 1930.08 to 1990.08 MHz. Substituting into equations
(1), (2), (3) and (4) above gives desirable first intermediate frequencies
of 192 MHz, 64 MHz, 338.52 MHz and 169.26 MHz respectively. The
64 MHZ IF is too low to provide sufficient image rejection when
operating over the 60 MHz wide 1900 MHz receive band. The 338.52
MHz IF is difficult to choose because of the unavailability of SAW
or crystal filters with a 30 KHz bandwidth at that frequency. The
solution of equation (1) or equation (4) is therefore preferred.
All the above solutions in tables 1-3 were for 1st LO high at 800
MHz and low at 1900 MHz, i.e. for the solution of equation (2).
These can be employed providing the range of the local oscillator
(51) is band-switched between 800 MHz and 1900 MHz operation. It
can be undesirable to attempt to cover in one band the entire tuning
range that would be required for operation at both 800 and 1900
MHz.
A search for solutions for the case of equation (1) yielded the
following result with the closest first IF to 192 MHz:
______________________________________ 1st IF 189.96 MHz 2nd Lo
190.08 MHz = 88/9 .times. the 19.44 (N1 = 88, M1 = 9) MHz crystal
TX offset 234.96 MHz = 89/72 .times. 2nd LO (N2 = 89, M2 = 72) (800
MHz) TX offset 270.00 MHz = 125/88 .times. 2nd LO (N2 = 125, M2
= 88) (1900 MHz) TX offset comparison frequency AT 800 MHz = 2640
KHz (actually 5280 KHz for the arrangement of FIG. 7, with N2 =
89, M2 = 36) TX offset comparison frequency at 1900 MHz = 2160 KHz
2nd LO comparison frequency = 21600 KHz Possible lst LO fractional-N
moduli: 1,2,4,8,11,22, (800 MHz) 44 or 88 and: 1,2,3,4,6,8,9,12,18,24,36
or 72 (1900 MHz) ______________________________________
The first LO phase comparison frequency is for example 240 KHz
if a fractional-N modulus of 8 is selected for both bands.
Alternatively, a fractional-N modulus of 24 may be selected to
give 720 KHz phase comparison frequency at 1900 MHz, but the phase
comparison frequency will still be 240 KHz at 800 MHz. The tuning
step size at 800 MHz will be 10 KHz with the same modulus of 24,
or even 5 KHz for the arrangement of FIG. 7. This is finer than
the 30 KHz needed, but is acceptable. 240 KHz is an adequate comparison
frequency for 800 MHz operation, and the higher comparison rate
of 720 KHz is desirable for 1900 MHz operation where the oscillator
phase noise is double that at 800 MHz.
A solution in accordance with equation (4) assumes division of
the high-band first local oscillator frequency by 3 for 800 MHz
operation. In other words, the divider (51b) of FIG. 7 must be changed
from a divide-by-2 to a divide-by-3 circuit. It is also necessary
then to change divider 45b to a divide by 3 circuit, in order for
the transmit frequency steps at 800 MHz to be correct. This solution
is not investigated further here, as it is not preferred for a dual-band
IS136 cellular phone, and in any case is an obvious extension of
the disclosed methods.
The invention can be used for dual-band/dual-mode radio telephones
in which compatibility with AMPS and IS54 (DAMPS) is desired in
the 800 MHz band together with compatibility with the PCS1900 (GSM-based)
standard.
The problem to be solved is that a radio is normally designed for
D-AMPS operation based upon the use of a 19.44 MHz crystal as the
most convenient common multiple of the 24.3 KS/S transmission symbol
rate, the 30 KHz channel spacing and the 8 KS/S voice digitization.
On the other hand, a radio is normally designed for GSM, DCS1800
or PCS1900 operation based upon a 13 MHz crystal, which is the lowest
common multiple of the transmission bitrate of 270.833 KB/S (13
MHz/48), the channel spacing of 200 KHz (13 MHz/65) and the 8 KS/S
voice digitization rate. This makes it difficult to merely integrate
a radio of one design with a radio of the other design, due to the
increase in parts count. Therefore it is desired to find internal
frequency plans that allow components to be designed that can operate
from either crystal frequency, and as another objective it is desired
to find reference frequency distribution schemes that will allow
operation with the same crystal reference frequency in any of an
AMPS mode at 800 MHz, a D-AMPS mode at 800 or 1900 MHz or a PCS1900
mode at 1900 MHz.
FIG. 12 illustrates a solution using both a 13 MHz and a 19.44
MHz crystal connected to reference oscillator (21), only one of
which however is activated at a time via a "select crystal"
control signal from digital logic (20).
A single intermediate frequency amplifier chip comprise a dual-crystal
reference oscillator (21), second LO and its control PLL (17) and
a dual-bandwidth second IF amplifier and second mixer (16). The
reference oscillator operates in one mode at 13 MHz and the second
LO is then controlled to 12.times.13 MHz. Alternatively, in a second
mode, the reference oscillator operates at 19.44 MHz and the second
LO is controlled to, for example, 155.52 MHz, which is sufficiently
close to 156 MHz that the same oscillator can be used, while also
being a multiple of 19.44 MHz (eight times 19.44 MHz).
The IF amplifier chip receives a downconverted signal from front
end chip (12) which is filtered either using Wideband IF filter
(15 WB) or Narrowband IF filter (15 NB). The filter center frequency
in the wideband mode is 150 MHz, which mixes with the 2nd LO of
156 MHz in that mode to produce a second IF of 6 MHz, which is fed
to digital signal processing (20) along with the RSSI signal. The
narrowband first IF filter operates at a center frequency of 120
KHz higher than the second LO of 155.52, that is at 155.64 MHz,
giving a second IF in the narrowband mode of 120 KHz, which is then
fed to the signal processing chip (20). The second IF signal at
either 120 KHz in the narrowband mode or 6 MHz in the wideband mode
is preferably further filtered in IF amplifier (16) using second
IF filters (not shown). In one implementation, the 120 KHz second
IF filters are integrated active bandpass filters having a passbandwidth
of approximately 30 KHz, and are fabricated as part of IF amplifier
chip (16,17,21). The 6 MHz 2nd IF filtering is performed by external
ceramic filters (not shown) of approximately 170 KHz bandwidth,
as used for TV sound IF stages.
When operating in the narrowband AMPS mode at 800 MHz, the duplex
spacing is 45 MHz and so the transmitter frequency is 45+155.64
MHz below the first LO. The TX offset would therefore be 200.64
MHz. However, as shown in FIG. 13, the TX mixer (46) mixes transmit
and receiver oscillators (45,51) at double the 800 MHz frequency,
and so produces an offset of 401.28 MHz. This has a highest common
factor of 1920 KHz with the second LO of 155.52 MHz, so divider
(47) divides the TX offset from TX mixer (46) by a first integer
N2 to obtain a first 1920 KHz signal, and divider (41) divides the
second LO from IF chip (30) by and integer M2=81 to produce a second
1920 KHz signal. The two 1920 KHz signals are compared in transmit
phase comparator (43) to produce an error signal. The error signal
is filtered and integrated in loop filter (44) to produce a control
signal for TX oscillator (45) to maintain it at the desired frequency,
which, when halved in divider (45b), is the desired 800 MHz transmit
frequency.
This frequency plan at 800 MHz may also be used for the D-AMPS
mode in the 800 MHz band. For operating in the D-AMPS mode at 1900
MHz, the duplex spacing is 80.04 MHz, so that the transmit offset
is 80.04+155.64 MHz=235.68 MHz. This is not simply related to second
LO frequency of 155.52 MHz; however, since only time-duplex modes
are used at 1900 MHz such that transmission and reception occur
in different timeslots and not simultaneously, the first local oscillator
may be sidestepped by the relatively small amount of 240 KHz between
transmit and receive so that a TX offset of 235.44 MHz may be used
instead of 235.68 MHz.
The slightly modified TX offset of 235.44 MHz shares a common factor
of 2160 KHz with the second LO of 155.52 MHZ. Thus in the 1900 MHz
D-AMPS mode, divider (47) divides by an integer N2 reprogrammed
to divide 235.44 MHz to 2160 KHz, while divider (41) is reprogrammed
to divide by an M2 of 72 to obtain 2160 KHz, the phase comparator
(43) now comparing signals at 2160 KHz instead of 1920 KHz.
Finally, to obtain the PCS1900 mode, where the duplex offset is
80 MHz, the transmit offset is 80+150 MHz, as the first IF is 150
MHz in that mode. The 230 MHz TX offset shares a common divisor
of 2 MHz with the second LO that is now 156 MHz. This mode is also
time duplex, and the first LO could be sidestepped to modify the
TX offset from 230 MHz to for example 234 MHz, which has the much
larger common factor of 78 MHz with the 2nd LO of 156 MHz. Nevertheless,
it may be advantageous to keep a phase comparison frequency of 2
MHz, which makes all phase comparison frequencies (1920, 2160 and
2000 KHz) sufficiently close to facilitate the use of a common loop
filter (44) and phase comparator (43). Otherwise, if desired to
take advantage of a larger common factor such as 78 MHz, a different
loop filter and even phase comparator may become necessary to provide
the desired closed loop characteristics of stability and lock-in
time. Thus the arrangement of FIG. 13 has deliberately aimed to
maintain roughly the same TX offset loop operational characteristics
of loop-bandwidth and lock-in time in all bands and modes.
The dual-mode, dual-band transmitter-receiver of FIGS. 12 and 13
assumes a double superheterodyne receiver is used in all modes.
In the narrowband AMPS and D-AMPS modes, the second intermediate
frequency is 120 KHz and the second IF filters are integrated, on-chip,
active filters; in the wideband PCS1900 modes, which can include
all GSM voice and data modes, satellite communication modes and
GPRS packet data modes, the second IF is 6 MHz, and the second IF
filters are more difficult to integrate at that frequency. An alternative
receiver architecture for the wideband mode is shown in FIG. 14,
in which the second IF in the wideband mode is zero frequency, otherwise
known as an "IF Homodyne", as opposed to an RF Homodyne,
which converts directly from the frequency received at the antenna
to zero frequency in one conversion step. The receiver of FIG. 14
converts from the frequency received at the antenna to zero frequency
in two steps, the first step converting to a first Intermediate
Frequency of 156 MHz and the second step converting from 156 MHz
to zero frequency by mixing with the 156 MHz local oscillator. Since
the first IF in FIG. 14 is now 156 MHz as opposed to the 150 MHz
of FIGS. 12 and 13, the TX offset for 1900 MHz is now 156+80=236
MHz, which still shares a common factor of 2 MHz with the 156 MHz
local oscillator. Thus the only change to FIG. 13 is that the value
of N2 for PCS1900 operation would change from 230/2=115 to 236/2=118.
If desired, a higher common factor of 4 MHz could be used by changing
N2 to 236/4=59 and M2 from 78 to 39, and M3.from 2 to 4 (or alternatively
changing the fractional-N modulus of N3 to accept a higher reference
frequency for third phase comparator (49).
The implementations of FIGS. 12, 13 and 14 use two different reference
crystals, although only one is active at any time. Nevertheless
this adds the complication that both crystals must be independently
temperature compensated, as every crystal has different individual
temperature compensation needs. Temperature compensation is carried
out by a "self-learning" technique, whereby the receiver
locks to a base station signal and then uses the base station signal
frequency as a basis for correcting crystal error. The prevailing
temperature is measured using a thermistor, and the correction applied
to the crystal is stored in a table against the prevailing temperature
in microprocessor memory in digital signal processor (20).
To simplify the temperature compensation as well as eliminating
the cost and board area associated with a second crystal, it is
therefore of interest to consider the solutions of FIGS. 15 and
16 using a single crystal. The solution of FIG. 15 is to choose
a compromise crystal frequency of 19.5 MHz. This is 1.5 times the
13 MHz from which PCS1900 bitrates are derived, and the bitrate
is still derivable as 19.5 MHz/72 as opposed to 13 MHz/48. 19.5
MHz is also close to the 19.44 MHz needed for D-AMPS modes, from
which the symbol rate of 24.3 KS/S is derived by dividing by 800.
When 19.5 MHz is used, the error is 0.3%, which would cause a timing
drift in the transmitted symbol stream of exactly half a symbol
period during transmission of a TDMA burst of 6.667 mS or 162 symbols
duration. In principle, such an error is no greater than must in
any case be anticipated by the receiver due to multipath propagation
causing transmission path delay variations of up to one symbol.
Nevertheless it is desirable to correct the transmitted signal so
that its errors do not compound the imperfections introduced by
the propagation path. To a first approximation, the symbol rate
error can be reduced by dividing the crystal frequency by 802 to
obtain the symbol rate with a residual error of 0.0585%, giving
a timing drift of less than one tenth of a symbol over a 162-symbol
burst duration. A further refinement can be made by means of a skip-counter,
which divides sometimes by 802 and sometimes by 803 in order to
create a more accurate approximation to the 24.3 KS/S symbol rate.
However, in one implementation, the 24.3 KS/S modulation is generated
digitally at the rate of 8 samples per bit. Several samples per
bit are used to represent the curved waveform of a symbol stream
that has been filtered using a root-raised-cosine filter frequency
response. Thus it is really desired to create an accurate approximation
to 8 times the symbol rate, or 194.4 kilosamples per second, by
dividing the crystal frequency sometimes by 100 and sometimes by
101. The number of times N1 division by 100 occurs and the number
of times N2 that division by 101 occurs will now be derived.
The D-AMPS frame repetition period of 20 mS represents 390,000
cycles of a 19.5 MHz clock as opposed to 388,800 cycles of a 19.44
MHz clock.
A timing generator is thus programmed to divide by 390,000 when
a 19.5 MHz clock is used as opposed to 388,800 when a 19.44 MHz
clock is used, in order to create the 20 mS repetition period. The
D-AMPS TDMA frame is divided into 3 timeslots, and one timeslot
is thus 130,000 cycles of a 19.5 MHz clock in duration as opposed
to 129,600 cycles of a 19.44 MHz clock. The first equation for N1
and N2 is therefore that
In addition, the total number of 1/8th symbol sample periods to
be created is 8.times.162=1296 as before, so the second equation
for N1 and N2 is
Solving these equations gives N2=400, N1=896.
Thus a skip counter is programmed to divide by 100 a total of 896
times interspersed with dividing by 101 a total of 400 times, creating
a total of 1296 1/8th symbol periods with no timing error greater
than of the order of half a clock period of the 19.5 MHz clock,
or 25 nanoseconds. FIG. 20 shows a skip counter design accomplishing
the above. A divider (100) is configured to divide either by 100
or by 101 according to a control input from accumulator (101), so
that successive output pulses from the divider (100) will be spaced
by either 100 cycles of the 19.5 MHz clock or by 101 cycles. The
accumulator (101) is configured as a modulo-81 accumulator, which
means that if after adding an increment, the value in the accumulator
is equal to or greater than 81, then 81 is subtracted from the accumulator
value and an overflow or carry pulse is generated. The carry pulse
output from accumulator (101) is used to cause divider (100) to
divide by 101.
If no carry is generated by accumulator (101) upon being caused
to increment by the last divider (100) output pulse, then divider
(100) counts 100 cycles of the 19.5 MHZ clock input to produce the
next output sample rate pulse. Else, if upon the last divider output
pulse causing the accumulator to increment and overflow, then the
accumulator carry output fed back to divider (100) causes the divider
to count 101 cycles of the 19.5 MHz clock input before producing
the next divider output sample rate pulse.
By setting the accumulator increment equal to 25, the accumulator
produces a carry pulse 25/81ths of the time, which is equal to 400/1296ths
of the time, this being the proportion of divide-by-101's calculated
above needed to produce the exact number 1296 of 8.times.symbol
rate pulses in a D-AMPS timeslot.
FIG. 16 shows the internal frequency plan using a 19.5 MHz crystal.
The first IF in D-AMPS mode is changed to 154.32 MHz to give high
phase comparator frequencies of 1320 KHz and 1080 KHz for transmit
phase comparator (43) in 800 and 1900 MHz operation respectively,
while also giving a high comparison frequency of 780 KHz at 2nd
LO phase comparator (31).
Yet another implementation of the invention is shown in FIG. 17,
this time using a 13 MHz crystal to derive all radio oscillator
frequencies, and a 19.44 MHz crystal connected to digital chip (20)
only to derive bit and digital sampling rates for the AMPS and D-AMPS
modes. The frequency plan for this case is shown in FIG. 18, in
which substantially the only difference from FIG. 16 is that the
second LO phase comparator now operates at 520 KHz.
In both FIG. 16 and FIG. 18, the main receiver synthesizer (the
first LO) operates as a fractional-N synthesizer with a modulus
of 5 (optionally 10 or 20) in PCS1900 mode, and 12 in AMPS and D-AMPS
modes.
It is possible to eliminate the 19.44 MHz crystal used only for
generating digital clocks by using the arrangement of FIG. 19, in
which the digital chip (20) creates its own 19.44 MHz clock by means
of an internal PLL when needed. To facilitate this, dividers (41)
and (42) are split into two dividers (41a,41b) and (42a,42b). Divider
41a divides the 2ND LO frequency of 154.44 MHz by 117 in D-AMPS
mode at 800 MHz to obtain 1320 KHz at which the transmit phase comparator
(43) operates. Selector switch 41c is operated to select the output
of divider 41a in this mode. During-this mode, divider 41b operates
simultaneously, dividing by 11 to provide a 14.040 MHZ output to
the digital chip (20). This frequency shares a common factor of
1080 KHz with the 19.44 MHz generated on the digital chip (20) by
means of a local PLL when needed. Divider (42a) operates at this
time to divide the operation frequency of phase comparator (43)
by a further factor of 2 to obtain 660 KHz, which is used along
with a modulus-11 fractional-N divider (43) to provide 60 KHz steps
of oscillator (51), which provides 30 KHz steps after division by
2 for 800 MHz AMPS or D-AMPS operation. For 1900 MHz D-AMPS operation
switch 41c selects instead the output of divider (42b) which is
14.04 MHz divided by 13, that is 1080 KHz. This is the desired frequency
to provide the duplex offset of 80.04 MHz at 1900 MHz compared to
45 MHz at 800 MHz. The selected 1080 KHz for phase comparator (43)
is then further divided by 3 in divider (42a) reprogrammed to divide
by 3, giving a 360 KHz phase comparison frequency for phase comparator
(49), which, together with the use now of a fractional-N modulus
of 12 for divider (48), gives 30 KHz tuning steps for oscillator
(51) during D-AMPS operation at 1900 MHz. For PCS1900 operation,
divider (41b) is programmed to divide by 12, dividing the second
LO of now 156 MHz to provide a 13 MHz clock output to the digital
chip (20). The 13 MHz is divided by 13 in divider (42b) to 1 MHz,
which is the operating frequency of phase detectors (43,48). Divider
(42a) is thus programmed to make M3=1. Using a fractional-N modulus
of 5 for divider (48) provides the desired 200 KHz steps in this
mode.
It has thus been shown above that the invention permits the construction
of dual-band, dual-mode transceivers using either a single crystal
reference or two crystals in a variety of ways to derive alternative
symbol rates of 270.833 KB/S and 24.3 KS/S, alternative channel
spacings of 30 KHz or 200 KHz, and transmit-receive duplex spacings
of 45 MHz, 80.04 MHz or 80.00 MHz.
Moreover, this flexibility is achieved with an improved architecture
as compared with the prior art, allowing the radio hardware to be
reduced to essentially three integrated circuit chips having a reduced
number of RF interconnections there between, thus minimizing risks
of internal interference and reducing power consumption.
An alternative to using a transmit offset frequency to control
generation of the transmit frequency, as described above, is using
a lock frequency. FIG. 21 illustrates a dual band transceiver employing
a lock frequency according to a second embodiment of the present
invention.
The dual band transceiver according to the second embodiment includes
similar components as the dual band transceiver described above,
e.g., a duplexor, a receiver front end, a receiver IF, modulators,
and a power amplifier. Although not shown, the dual band transceiver
according to the second embodiment also comprises DSP means (20)
as described above.
For illustrative purposes, the second embodiment is described with
reference to operation according to the PCS and AMPS/DAMPS standards.
It will be appreciated, however, that the invention is also applicable
to other standards.
Referring to FIG. 21, for frequency duplex operation, the Duplexor
(11a) includes a Diplexer or Switch, PCS and AMPS/DAMPS Couplers,
a PCS Duplexer or RX/TX switch, and an AMPS/DAMPS Duplexer. A signal
received by the antenna 10 is diplexed or switched, to select the
band of operation. The PCS and AMPS/DAMPS Couplers detect the power
level of the system and communicate with the DSP (20) which ensures
that the system is operating at the appropriate power level for
the band of interest. The PCS Duplexor or RX/TX Switch and the AMPS/DAMPS
Duplexor couple the appropriate input to the receiver front end
(12a).
The duplexors can be replaced with a switch or used in conjunction
with an isolator or circulator, depending on the requirements for
a particular band. Similarly, the switch may be replaced with a
duplexer or used in conjunction with an isolator or circulator.
The receiver front end (12a) comprises one or more low-noise amplifiers
and downconverting mixers. The receiver front end (12a) may contain
separate amplifiers and mixers for each band, as shown in FIG. 21.
Filters for each band may be included in the receiver front end
(12a) or implemented as separate components, as shown in FIG. 21.
A 1st LO (51) and a frequency doubler (59) may also be included
in the front end (12a) or implemented as separate components as
shown in FIG. 21.
For low band operation, the 1st LO frequency from the 1st LO (51)
is delivered to a mixer in the receiver front end (12a). For high
band operation, the 1st LO frequency is doubled in a doubler (59),
then delivered to a mixer in the receiver front end (12a). The 1st
LO frequency or the doubled 1st LO frequency, depending on the band,
is mixed with the desired receive frequency signal to produce a
first intermediate frequency signal which is filtered in the IF
filter (15). The desired receive frequency is selected by tuning
the 1st LO (51) to a frequency equal to the sum or the difference
of the desired receive frequency and the first IF, using the 1st
LO PLL (14). The 1st LO (51) is a channel stepper which is tuned
by the 1st LO PLL (14) to an integer multiple of a basic tuning
step size, derived from a reference frequency source, e.g., a crystal
frequency oscillator (21,22). The 1st LO (51) operates at a frequency
adapted to the lower of the two-possible receive frequency bands
to convert received signals to the desired first IF. For high band
operation, the 1st LO PLL (14) tunes the 1st LO (51) to a doubled
integer multiple of a basic tuning step size, using the frequency
doubler (59).
The 1st LO PLL (14) compares the 1st LO frequency or the doubled
1st LO frequency to the reference frequency and generates an error
signal which controls the 1st LO (51). The 1st LO PLL (14) may include
a switch to select whether to compare the 1st LO frequency or the
doubled 1st LO frequency to the reference frequency. If the lower
frequency is always selected, this switch may be eliminated. By
selecting the LO frequency for comparison in the 1st LO PLL (14),
depending on the band, the same channel stepping (e.g., 30 KHz)
can be maintained for both bands. In addition, current is saved
by not operating the 1st LO PLL (14) at twice the low band frequency
when low band operation is desired.
The receiver performs a second frequency down conversion in a receiver
IF circuit (16a) which comprises, e.g., a mixer for mixing the first
intermediate frequency down with a 2nd LO frequency from the 2nd
LO (33). Although shown as separate components, the 2nd LO (33)
can be contained in the receiver IF circuit (16a). The 2nd LO (33)
is controlled to the desired frequency by the IF PLL circuit (17)
which compares the 2nd LO frequency with a reference frequency from,
e.g., a crystal oscillator (21) and produces an error signal.
The transmitter comprises a transmit frequency oscillator (45)
for generating the transmit frequency. The transmit frequency is
offset from the 1st LO frequency by the duplex spacing, combined
with the first intermediate frequency. In the past, the transmitter
frequency offset has been used to control the generation of the
transmit frequency. A problem with using offset in this manner is
that up-conversion products are generated which have to be filtered.
This adds a filtering burden to the transmitter.
According to the second embodiment, this problem is avoided by
mixing the 1st LO frequency down with the transmit frequency to
produce a lock frequency and comparing the lock frequency and the
reference frequency in the TX PLL (19). The comparison result is
used to control the generation of the transmit frequency by the
transmit frequency oscillator (45).
According to an exemplary embodiment, the transmit frequency signal
generated by the transmit frequency oscillator (45) is multiplied
by an integer X1, X2, . . . or XN in a buffer/multiplier (53) to
produce an "on-channel" transmit frequency. This frequency
is then mixed in a TX mixer (46) with a 1st LO frequency from 1st
LO (51) to produce a lock frequency. The lock frequency is compared
with a reference frequency from e.g., crystal oscillator (21) or
any other suitable frequency source such as the second LO (33),
and the result is used to control the TX LO (45).
A frequency doubler (54) doubles the low band transmit frequency
to produce a high band transmit frequency when operation in the
higher of the two frequency bands is desired. The high band transmit
frequency and the low band transmit frequency signals are amplified
in amplifiers (55) and (56), respectively, then modulated in the
I/Q modulator (18) with information from, e.g., the DSP (20).
The I/Q information is directly modulated onto the transmitter
carrier via the I/Q modulator (18). The output of the doubler (54)
is modulated when operation at the high band is desired, while the
output of the multiplier (53) is modulated for low band operation.
The modulated signal is amplified in variable gain amplifiers (57)
and (58) which may be implemented as a single, wideband device or
as individual narrowband devices, optimized for each band, as shown
in FIG. 21. The output of the variable gain amplifiers (57) and
(58) is amplified in a transmit power level using power amplifier
(13), which may be implemented with a narrowband power amplifier
as shown in FIG. 21 or with a broadband power amplifier. The mixer
(46), multipliers (53) and (54), amplifiers (55) and (56), modulator
(18), and variable gain amplifiers (57) and (58) may be implemented
as a single device, as shown in FIG. 21.
The dual band radio architecture according to the second embodiment
can be modified in several ways, depending on the particular standard
in each band which is to be implemented. For example, the on-channel
TX LO (45) may include a band switch if that is helpful in the tuning
range between band use of the oscillator.
In addition, the output of the TX mixer (46) may be filtered in
a filter (46a) before comparison in the TX PLL (19), as shown in
FIG. 22. This reduces spur content.
The I/Q modulator (18) may be broadband, covering both transmit
bands, as shown in FIGS. 21 and 22. Alternately, the I/Q modulator
may be narrowband, optimized for each band, as shown in FIG. 23.
As shown in FIG. 24, the transmit frequency may be delivered directly
to the TX mixer (46) from the TX LO (45). In this case, the multiplier
(54) may be replaced with a buffer (54b), and a divider (53c) may
be added to the low band path.
As shown in FIG. 25, the transmit oscillator frequency may be at
any integer multiplier/quotient of the desired transmit signal,
including the integer 1. For this purpose, multiplexers/dividers
(53a-b) and (54a) are included for the low band and the high band,
respectively. A switch may be included in the high band path, to
select multiplication/division in the multiplexer/divider (54a),
depending on the multiplication/division in the multipliers/dividers
(53a) and (53b). It will be appreciated that the transmit frequency
may be delivered to the TX mixer (46) from any multiplier/divider.
Also, as shown in FIGS. 25 and 26, variable gain amplifiers (59a)
and (59b) may exist at the I/Q modulator input, in addition to or
instead of the variable gain amplifiers (57) and (58) at the modulator
output.
The dual band transceiver according to the second embodiment has
many advantages. For example, the transmitter employs an on-channel
transmit frequency to generate a lock frequency, as opposed to using
an offset signal to produce the transmitter carrier. Consequently,
fewer spurious signals are generated. An on-channel arrangement
also reduces the filtering requirement to satisfy the transmitter
masks of radio standards. This is especially helpful for radios
that are not full duplex, such as those employing TDMA standards,
which do not inherently demand significant transmitter band filtering.
Surface acoustic wave (SAW) filters are customarily added to remove
spurs generated in the conversion process that exceed the transmit
mask.
Another advantage is that the lock frequency is the same type frequency
that an offset oscillator would produce in a transmitter offset
approach. Because this signal is a mix down product, it can be at
a significantly lower power level with lower harmonic energy than
what is generated by an oscillator circuit. With lower levels, there
is less danger from signals leaking around filters and getting out
of the transmitter directly or through mixers. Thus, this embodiment
has the benefit of reduced levels of spur signals.
The TX PLL (19) according to the second embodiment, performs a
strong filtering function acting as a low pass filter with a cutoff
frequency anywhere from 1 KHz to 1 MHz. The VHF frequency that is
used as a reference frequency is filtered to a very small value
before it is inserted into the transmitter up-conversion process.
Therefore, very low spurious content is maintained. Filtering of
a 50 to 250 MHz signal is much easier at 1 MHz than at 1 and 2 GHz.
Thus, this embodiment has the benefit of easier filtering of generated
spur signals in the transmitter.
Another benefit is that current is saved in the transmitter in
comparison to the offset approach, because the number of mixer/modulator
stages between the IQ modulation and the transmit output frequency
is reduced to one. For linear transmitters, every linear mixer adds
current. In addition, current is saved by activating the doubler
(54) only when operating in the high frequency band. Current is
also saved by selecting the low band (900 MHz) input to the main
receiver LO synthesizer to avoid running the synthesizer at twice
the frequency when operating in the low band.
Another benefit is that a single channel stepper is shared for
both the transmitter and the receiver channels in both bands, reducing
space, current, and costs. Also, one TX LO and one 1st LO are shared
for both bands, thus further reducing space and costs.
Yet another benefit is that full duplex operation is allowed for
both bands. Generally, in a dual band transceiver, the duplex spacing
is different between bands. For a shared channel stepper, this can
be a problem if the channel stepper has to make up the difference
in duplex spacing between bands. According to the second embodiment,
the transmit loop resolves the duplex spacing difference, instead
of the channel stepper, thereby enabling full duplex operation in
both bands.
The dual band transceiver according to the second embodiment also
enables the RX PLL (14) to step in full channel spacing for both
the low and high bands. This avoids incrementing the TX PLL (14)
in one-half channel spacing (15 KHz for 30 KHz channel spacing required
in IS-136, IS-137) for the 1900 MHz band in a 900/1900 MHz dual
band hand set.
In addition, the invention provides flexibility in the TX PLL (19).
The transmit frequency provided to the TX mixer (46) from the transmitter
can be at the lower band always or switched between bands.
Although the description of the embodiment above is directed to
a dual band phone, it will be appreciated that it is also applicable
to a single band phone by removing the components corresponding
to the second band or by simply using only the components for one
band.
The invention may be useful in contexts other than cellular radio.
The invention may be adapted by a person skilled in the art using
the above teachings while remaining within the spirit and scope
of the invention as described by the following claims. |